反激的典型波形

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1、 .wd.反激变换器的例子Analysis of basic waveforms 根本波形分析The analysis of the basic waveforms will be done on a simulated example of a flyback converter operating in discontinuous conduction mode. Typical drain-source voltage waveform of the primary side switch is shown in Fig. 16.在电感电流断续模式下运行的反激变换器的典型一次侧漏源极开关

2、电压波形见图。Fig. 16 Typical drain-source voltage of the MOSFET in a flyback图反激变换器的典型漏源极电压These drain-source voltage waveforms can be theoretically distinguished into typical elements. Different physical phenomena influence the waveform at given time interval. Fig. 17 and Tab. 4 demonstrate the main eleme

3、nts of the voltage waveform. The superposition of all these elements results in a typical drain-source voltage shown in Fig. 16.这些漏源极电压波形能用典型的理论来描述。各个时间段有不同物理现象影响这些波形。图和平台描述了电压波形的主要原理。把这些原理按时序整合呈现出图所示的典型漏源极电压。Fig. 17 Main elements of the drain-source voltage图漏源极电压的主要原理原理：开通期间的电压下降过程原理：在开通期间因寄生震荡产生的电

4、流尖刺原理：关断期间的电压上升原理：缓冲电路的钳位电压原理：钳位过程完毕后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡原理：磁芯存储磁能释放完毕后主要由场效应晶体管输出电容和变压器电感引起的寄生振荡原理：反激变换器释放磁能期间的反射电压原理：与直流母线电压等幅的主要方波Tab. 4 Main elements of the drain-source voltage平台漏源极电压的主要原理The spectrum of the whole drain-source waveform (Fig. 16) is presented in Fig. 18.图所示的漏源极电压呈现的电磁干扰频谱见

5、图。Fig. 18 Spectrum of the drain-source voltage (as shown in Fig. 16)图图所示的漏源极电压呈现的电磁干扰频谱The spectra of the main elements of the drain-source voltage can be found in Fig. 20. Fig. 19 is exactly the same as Fig. 17 and has been repeated here for better under-standing.图描述了漏源极电压主要原理产生的电磁干扰频谱。为便于理解，将图映射成图

6、。Fig. 19 Main elements of the drain-source voltage (repeated, same as Fig. 17)图漏源极电压的主要原理正确重复图Fig. 20 Spectra of the main elements of the drain-source voltage图漏源极电压主要原理产生的电磁干扰频谱This method allows associating certain parts of the spectrum with their root causes, i.e. the peak at 20 MHz in the spectru

7、m of the drain-source voltage is caused by the parasitic oscillation due to the output capacitance of the MOSFET and the leakage inductance of the transformer.这种方法可以确定电磁干扰频谱中某些频点的来源，也就是说漏源极电压产生的电磁干扰频谱中的兆赫兹峰点是钳位过程完毕后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡产生的。The analysis of the drain current of the primary switch

8、 will be done in the same way. Fig. 21 demonstrates a typical drain current in a DCM flyback.对一次侧开关的漏极电流进展分析采用一样的方法。图展示出一个工作于电感电流断续模式反激变换器的典型漏极电流。Fig. 21 Typical drain current in a flyback图反激变换器的典型漏极电流This waveform can be presented as a superposition of the following elements (Fig. 22 and Tab. 5). T

9、he superposition of all these elements results in a typical drain current shown in Fig. 21.这个波形可以被看作是以下原理的叠加图和平台。全部这些波形的叠加整合结果变成图所示的典型漏极电流。Fig. 22 Main elements of the drain current图漏极电流的主要原理原理：漏极电流的主要三角波形原理：在开关开通期间因寄生分布电容引起的电流尖刺原理：钳位过程完毕后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡原理：磁芯存储磁能释放完毕后主要由场效应晶体管输出电容和变压器电感引起

10、的寄生振荡Tab. 5 Main elements of the drain current平台漏极电流的主要原理The spectrum of the whole drain current waveform (Fig. 21) is presented in Fig. 23.全部漏极电流波形产生的电磁干扰频谱图呈现在图。Fig. 23 Spectrum of the drain current (as shown in Fig. 22)图漏极电流产生的电磁干扰频谱与图一样The spectra of the main elements of the drain current can be

11、 found in Fig. 25. Fig. 24 is exactly the same as Fig. 22 and has been repeated for better understanding.漏极电流主要原理产生的电磁干扰频谱见图。图和图一样。Fig. 24 Main elements of the drain current图漏极电流的主要原理Fig. 25 Spectra of the main elements of the drain current图漏极电流主要原理产生的电磁干扰频谱As in case of drain-source voltage this me

12、thod allows to associate the elements of the drain current waveform with its contribution to the whole spectrum. For example, the peak at 20 MHz in the spectrum is caused by the parasitic oscillation due to the output capacitance of the MOSFET and the leakage inductance of the transformer.就象漏源极电压的例子

13、那样，用这种方法也可以找出漏极电流的哪一局部对电磁干扰频谱产生影响。举例说明，兆赫兹的峰点是钳位过程完毕后主要由场效应晶体管输出电容和变压器漏感引起的寄生振荡产生的。This method of separating the waveform in time domain into its main elements helps to find out what part of the spectrum in frequency domain caused by what related physical phenomena. The separation into main elements

14、 should be done in respect of reasonable events in the power circuit like on and off slopes, oscillations, clamping, snubbering, reflected voltage, etc.这种在时域里对主要原理进展拆分的方法有助于找出产生电磁干扰频段的干扰源。这种离析主要原理的手法有助于合理审视电源电路里诸如变化速率、振荡、钳位、缓冲、反射电压等过程。In this flyback example only the primary switch has been analyzed

15、 as active source of electrical noise. There are also others, like secondary side diodes or synchronous rectifier, control IC (especially its gate drive), etc. In order to obtain more complete analysis all these interference sources have to be analyzed.在这个反激变换器里只对一次侧开关进展电磁噪声产生的分析。但是还有其他的局部，象二次侧的二极管或

16、同步整流器、控制集成电路尤其是它们的栅极驱动等等。按顺序分析将获得更完善的关于这些电磁干扰源的解析。However, it is impossible to predict the conducted EMI spectrum using this approach due to the fact, that only interference sources are considered. There is no analysis of the spreading paths of the interference in this method.然而，这种方法不可能预知用频谱反映的电磁干扰的

17、实际行为，仅仅是干扰源被重视起来。在那里没有对分布参数产生的干扰进展分析的方法。Nevertheless, the association of harmonics root cause with the respected physical phenomena will reduce the efforts of EMI reduction. The impact of the identified root cause can be reduced not only by filtering, but also by means of influencing the root cause

18、itself.不过，重视物理现象并不能成就电磁干扰的降低。降低干扰并不仅仅是滤波，也同样意味着干扰源自身的影响。Operation modes of discontinuous flyback converter 电感电流断续工作反激式变换器的运行模式The flyback converter running in discontinuous conduction mode can be operated in hard switching or quasi resonant (or valley switching, or ZVS) mode regarding the primary si

19、de switch. The difference between a hard switching and quasi resonant flyback converter is the turn on time point of the primary switch. In a hard switching mode the turning on of the MOSFET is not synchronized with the drain-source voltage value. This type of converters runs mainly in fixed frequen

20、cy mode.电感电流断续工作的反激式变换器一次侧开关可工作于硬开关或准谐振或谷值开关或零电压开关模式。硬开关和准谐振反激变换器之间的差异在于一次侧开关的开启时间点。在硬开关里场效应晶体管的开启波形拐点并不和漏源极电压值同步。这种变换器大体上运行于固定频率模式。In a quasi resonant mode the resonant circuit determined by the output capacity of the MOSFET and the inductance of the transformer will be utilized to switch on at low

21、est possible value of the drain-source voltage. This circuit starts to oscillate at the end of the current flow through the secondary side of the transformer, hence at the end of the flyback phase. The MOSFET will be turned on at the minimum of this oscillation. The quasi resonant approach uses this

22、 oscillation to achieve minimum voltage switching during turn on for the MOSFET. This operation mode runs at a variable frequency.在准谐振模式里，由变压器电感和场效应晶体管输出电容引起的谐振促使开关的开通时刻发生在漏源极电压的最小值上。这种电路在电流从变压器二次侧流尽以后反激回扫过程完毕开场振荡。场效应晶体管将在振荡幅值的最小值开启谷值开通。这种运行模式工作在可变的频率上。Higher amplitude of the oscillation results in

23、lower drain source voltage level at which the MOSFET turns on correspondingly lower switching losses and higher efficiency of the system.更高幅值的振荡导致场效应晶体管更低的漏源极开通电压幅值来产生更低的开关损耗和更高的系统效率。To achieve high oscillation peaks, the design of the transformer has to be set to high reflected voltage. This increa

24、se of the reflected voltage results in a higher drain-source voltage blocking MOSFET and longer duty cycles.要到达比拟高的振荡电压峰值，变压器的反射电压必须设置的比拟高。增加的反射电压导致使用更高漏源极击穿电压的场效应晶体管和更大的开关占空比。Comparison of three different flyback solutions has been made. All of them have been operation at 300 kHz, bus voltage of 40

25、0 V, output power of 120 W, output voltage of 16 V. These design included different modes of operation and different values of reflected voltage, resulting in different MOSFETs voltage ratings:比拟现有的三种反激变换器。它们都工作在千赫兹，直流母线电压伏特，输出功率瓦特，输出电压伏特。这些设计包含不同的运行模式和反射电压等级，因此使用不同电压等级的场效应晶体管：lHard switching flybac

26、k with CoolMOS 600V, reflected voltage of 100V l硬开关反激变换器使用伏特CoolMOS，伏特反射电压lQuasi resonant flyback with CoolMOS 600V, reflected voltage of 100V l准谐振反激变换器使用伏特CoolMOS，伏特反射电压lQuasi resonant flyback with CoolMOS 800V, reflected voltage of 390V l准谐振反激变换器使用8伏特CoolMOS，39伏特反射电压The clamping snubber circuit wa

27、s set to the rated breakdown voltage of the MOSFET (600 V and 800 V respectively).钳位缓冲电路被设定在场效应晶体管的额定击穿电压上分别为伏特和伏特。Flyback in hard switching mode with 600V MOSFET 使用伏特场效应晶体管的硬开关反激变换器The hard switching approach (as shown in Fig. 26) doesnt consider the minimum drain-source voltage. The MOSFET will be

28、 turned on hard, in this case at a voltage level of 500 V (at time point 3.3 s). The discharge of circuits parasitic capacitances leads to a high current spike during turning on.硬开关图所示几乎不考虑漏源极电压的最小值。场效应晶体管开通应力大，在这个例子里，开通电压在伏特在.微秒的时间点。由寄生电容引起的泄放电流在开通时产生很高的电流尖刺。Fig. 26 Drain-source voltage and drain c

29、urrent of hard switching 600V flyback图伏特硬开关反激变换器的漏源极电压和漏极电流Flyback in quasi resonant mode with 600 V MOSFET 使用伏特场效应晶体管的准谐振反激变换器The drain-source voltage (Fig. 27) starts oscillating at the end of the flyback phase and reaching the minimum of 300 V when the MOSFET turns on. 漏源极电压图在反射过程完毕后并减小到伏特时场效应晶体管

30、导通。The duty cycle is lower compared to an 800 V solution due to a lower reflected voltage of 100V. Shorter duty cycle for the same output power results in higher peak currents on the primary side.因为伏特的反射电压，比拟伏特解决方案它有更小的占空比。小占空比实现同样的功率输出必须使用更高的一次侧峰值电流。Fig. 27 Drain-source voltage and drain current of

31、 quasi resonant 600V flyback图伏特准谐振反激变换器的漏源极电压和漏极电流Flyback in quasi resonant mode with 800 V MOSFET 使用伏特场效应晶体管的准谐振反激变换器The drain-source voltage (Fig. 28) starts oscillating at the end of the flyback phase and reaching the minimum of 100V when the MOSFET turns on. The turning on current spike is low.

32、漏源极电压图在反射过程完毕后并减小到伏特时场效应晶体管导通。开通电流尖刺比拟低。The duty cycle is higher compared to a 600V solution due to a higher reflected voltage of 390V. Longer duty cycle for the same output power results in lower peak currents on the primary side.因为有伏特的反射电压，所以有比伏特解决方案更大的占空比。更大的占空比实现同样的输出功率可以使用更低的一次侧峰值电流。Fig. 28 Dra

33、in-source voltage and drain current of quasi resonant 800V flyback图伏特准谐振反激变换器的漏源极电压和漏极电流Comparison of spectra 干扰频谱比拟The spectra of the drain-source voltages for corresponding flyback design (Fig. 26Fig. 27 and Fig. 28) are shown in Fig. 29.相应设计的反激变换器图、图和图的漏源极电压干扰频谱如图所示。Fig. 29 Spectra of the drain-s

34、ource voltage (simulated)图漏源极电压的频谱仿真As it can be seen the voltage spectrum of the 800V quasi resonant flyback is higher at frequencies below 1 MHz, and is getting lower above 1 MHz compared to both 600V designs. This can be explained by two major differences of the 800V drain-source voltage waveform

35、. First, the clamping voltage during the MOSFETs turning off is 800V, what is higher then of 600V. It leads to higher harmonics amplitudes in lower frequency range. Second, the turn on occurs in voltage minimum due to quasi resonant switching, which results in lower spectrum in higher frequency rang

36、e.和伏特设计比拟，伏特准谐振反激变换器的电压频谱在兆赫兹以下更高一点，在兆赫兹以上开场变小。这里有两条理由可以解释伏特漏源极电压波形的两个差异。第一，在场效应晶体管关断期间钳位电压是伏特，高于伏特。它产生低频段的高振幅。第二，开通发生在准谐振开关电压的最小值，这导致更高频段频谱中更低的幅值。Due to the fact, that the 800V quasi resonant flyback has lower peak current, its spectrum is significantly lower across almost complete frequency range. 事实上，伏特准谐振反激变换器拥有更低的峰值电流，它的频谱意味着在全频带有更低的幅值。The 800V quasi resonant design with lower current peak and lower drain-source voltage during turning on of the MOSFET demonstrates advantages in conducted EMI spectra regarding the primary side.拥有更低峰值电流和场效应晶体管漏源极开通电压的伏特准谐振设计展示出一次侧传导电磁干扰降低的优势。